I’ve used the LR8 reg a few times here and there, always with satisfactory results. Yet, I’ve never spent time looking into it with more detail, always helped me to fix a voltage reference needed. Over time, I shifted towards the gas regulators and MOSFET-based voltage regulators where needed.
I also wrote extensively about the “sound” of regulators and why I’m reluctant (as many out there) to include closed loop regulators in my HT supplies. Having said that, having a tight closed loop reg to generate a reference and use a cap multiplier is a great alternative to provide a very clean supply.
The LR8 can do a great job here for voltages below 430V-ish if you marry him with a nice MOSFET like this:
LR8 regulator
The LR8 needs a 1.2V reference across the ADJ and output pins. R1 and Rtrim+R3 provide the adjusting network to dial the output voltage. See the datasheet for more info. R3 sets the minimum voltage and Rtrim provides the range needed for adjustment. Of course you can tweak this network for your needed voltage. I just simply worked this out to provide a 90V reference. RLPF and CLPF are the low pass network feeding the pass MOSFET. The LR8 has some BW noise like every reg which spreads into HF. Although there is no info on the datasheet, I remember measuring this but can’t find the data at hand. The 220K+47uF combination on the LPF network seem to provide a good balance on the rejection at 50Hz as well as the transition response to slowly rise the output in about 45sec.
The spice model I use isn’t trustworthy as is not official. It shows a notch which can be moved to about 100Hz with the above RC network values chosen. Proof is in the pudding and have to measure this to see if it’s somewhere near this simulation. Either way, the rejection is over 80dB across the band and if it measures as good as this, then we are all happy.
I’ll do a small PCB for this so if there are interested on it, just let me know.
After being out of action for over a month due to visits, holidays and business travel, I finally got the opportunity to get my hands back on the 600V bench supply. I need to repair my valve tracer, but firstly need the bench supply back again. Otherwise won’t be able to do all tests I want to around my 814 DC-coupled SE amplifier and many driver stages I want to try on my workbench before moving to the next stage.
An improved design to avoid fireworks
Tired of my old HT (+600V) variable bench power supply to suffer collateral damage when accidentally shorted whilst testing transmitting valves for output stage (i.e. FQP3n80c MOSFET passive regulator blowing out), I decided firstly to decide a simple and yet effective valve stabiliser. As nothing comes for free, these were my design constrain factors:
Input raw supply is +620V @ 100mA
Filament secondary winding is 15V @1.5A
No additional secondary winding is available for a floating screen supply (e.g. pass valve is pentode)
Output voltage ideally should be 0-600V
So with the restriction of not using a pentode as pass valve, I looked out for candidates to match my requirements and instantly thought about GU-50 in triode-strapped mode. Yes, I know that UG2 limit is 250V, not 1,000V as anode max voltage. But, in triode strapped specs are not shown. As recently checked this with the 814 triode strapped, and seems to be ok UG2=Ua in triode mode. 7N7 also said this was ok and Morgan Jones previously tested this as well with similar valves.
So, question here initially was: could the GU-50 withstand 600V in triode mode or should I needed to look out for other options?
After asking for some help in DIYaudio forum to see what was the best option on this topology and the recommendation was to use the GU-50 in “right-handed mode”:
GU50 right-handed
In this mode GU-50 was able to provide the regulation required (or close to it) with minimum driving requirements (i.e. 0V to +15V)
First version of the passive regulator:
The MOSFET is in source follower mode to provide the necessary grid current, albeit not sure how much grid current the GU-50 needed in right-handed mode. So the value of R6 must be adjusted on test. In order to survive an output short, R6 needs to be in the order of 100K and 3 or 4W to avoid blowing up the 18V zener protecting the grid.
The circuit above is very interesting as the anode dissipation of the GU-50 is very handy for this setup and requirements. Is important to highlight that the circuit has no regulation, so if this is a requirement a different topology must be explored.
If we look at the sandy option here, then the following equivalent is comparable:
R2 and R3 are required to protect silicon from output short. Q1 provides current limitation with R6. This topology suffers from same issues related to regulation as previous circuit.
A big issue on the two circuits presented before is that the potentiometer is stressed at full raw HT supply. This is far from ideal and despite the specific power requirements of the pot, we also need to ensure that the part can withstand the voltages used.
A slight modification (requiring an additional LT supply) can solve this problem:
Now P1 is connected to 15V. The gain of the M2 stage is significant so stability of circuit above is an issue now.
So if we have already introduced an LT supply in the circuit, then is a better choice to look at a feedback regulator.
A more complex circuit indeed, but a more effective one. The op-amp provides regulation by sensing output from R7 and R8 divider and comparing it to the stable reference from the output of P1. R10, R9 and C1 limit the HF gain response of the op-amp. R12/C3 and the 10pF FKP2 Wilma cap across R7 optimise the HF response of the overall regulator. D1 will protect the op-amp input (specially if 10pF cap is fitted). R1 is an additional protection for M1. M1 requires a good sink if wider regulation is needed. When output voltage is low and high current is drawn, then M1 is bearing all the effort and will dissipate a lot of heat (just do the maths).
The raw supply stays the same, however the gyrator stage was optimised as shown below:
The MOSFET was changed for am 1000V TO220FP plastic package one which is better from an insulation perspective given voltages used in this circuit. R5 and R6 changed to 150K 3W ones to provide protection to the zener and M1 in case output is shorted.
