Years ago when I built my analogue curve tracer I added a small, yet very effective valve leakage test circuit. Due to my laziness, I failed to test a transmitting tetrode which I bought on-line and despite being NOS it damaged my uTracer. I followed the repair and re-calibration process and got the tester back again running, however, I regretted not having used this simple one step test I normally did before. Lesson learned, now I do use it back again!
Here is the circuit in case you don’t have a proper tester and you want to build something similar yourself:
You can test for leakage current using a simple amplifier made out of a NPN transistor and an indicator. In this case I used a Russian Neon (80V/0.5mA) and the existing supply on my tester (+/-80V). You can replace all this with a simple LED and the supply you have at hand. The circuit is designed to turn on the bulb when 5 μA leakage current is provided on the base of Q1 thanks to grounding the valve element next to the one under test. So for example if we want to measure cathode to grid leakage, we simply ground the cathode and we connect the tester to the grid. Same process is repeated with the other valve elements.
When I asked for some help in the DIYAudio forum, someoone gently recommended this text. Unfortunately I don’t read German, but what I got out of this adivce was:
valves with poor vacuum (i.e. failed the test described on the procedure in 18A)
preamp valves with less than 4 μA are usable
output valves whit less than 10 μA are usable
Valves that are good and show little Gas on the gas test:
should have less than 0,6-1μA for preamp valves
And should have less than 1.5-2μA for output valves
So the 5μA threshold was good enough in my view. It does work well and the beauty is that when neon light is very dim is an indication that it may be a workable valve despite the tiny leakage in particular with output valves.
I’m a heavy user of CCS loads. I generally use them to test my valves regardless of using my curve tracer or not. I tried multiple CCS types in the past with good results until I ended up burning one FET or protection zener or whatever due to the abuse of it.
Testing high current loads is not easy at high voltages. The DN2540 is rated at 400V. Not enough. You can use an expensive 01N100D which is another depletion 1KV MOSFET that has a lower Ciss (54pF against 200pF) or you can look at the cheaper enhancement FETs which require a different bias arrangement. If we are looking at modifying the classic cascode self-bias pair, it is a convenient opportunity to improve the VDS bias of the lower FET to improve the frequency response by lowering the Ciss. Remember that in a FET the Ciss is proportional to the VDS. The classic cascode pair has a disadvantage as the lower FET is biased with VDS lower than 1-2V to ensure the upper FET is biased correctly.
For some time I’ve been postponing the conclusion of a half-finished project. This is one of the many projects that I have around as many of you, but it was time to complete it as just some minor bits were outstanding.
My interest in measuring valve transconductance was very high since my early days of involvement with hollow state technology. The old valve tester I acquired didn’t measure it, I tried many ways to measuring it with different methods until I settled with using a CCS load and an AC meter as described here.
The problem I found though was that my true RMS AC meter in low scale (i.e. 100mV AC) didn’t like a significant DC voltage drop across the sensing resistor. Not sure why, but either way I wasn’t happy either without decoupling the anode to the sensing part of the circuit when using high voltages.
After listening to a great incarnation of the 4P1L PSE in filament bias output stage from Andy Evans, I decided to have a look at the impact of unmatched pairs of triodes from a distortion point of view. Main reason was that when listening to Andy’s amplifier I noticed a bit of an uncomfortable treble with some strings. Perhaps the increase of odd harmonics, but wanted at least to see what was all about.
4P1L are very easy to match. you can easily get a pair with equal mu. Just randomly I picked from my collection a pair of valves with a difference of 0.5 in mu.:
4P1L PSE unmatched pair
THD is about 0.03% mainly driven by H2. It happened that one 4P1L from the pair had 0.02% where the other had nearly 0.04% distortion. The difference between H3 and H2 is about 8dB.
Then looked at a more closely matched pair (0.03 mu difference). The distortion wasn’t surprisingly different:
4P1L PSE matched pair
Again, nearly 0.03% and difference between H2 and H3 is down to 7.5dB.
Looking at the individual performance of the 4P1L, now biased at 30mA and similar anode voltage, we can see that despite having a lower THD, the difference between harmonics is just 5dB. This is the THD of the other 4P1L from the pair:
4P1L PSE test rig
Well, how rthis compares to a 2a3/6C4C? The latter valves are two triodes physically connected in parallel inside the same envelope. So, no matching can be done:
The previous was a low distortion 6C4C I have. Distortion is higher than 4P1L PSE, but not that much. H3 – H2 difference is about 12dB.
My early thoughts:
4P1L are very easy to match
4P1L PSE performs really well. Distortion of the pair is lower than a 6C4C performing at same level.
H3 component is higher in PSE and this could be the reason why is more noticeable when listening to strings – as I proved in practice.
After being out of action for over a month due to visits, holidays and business travel, I finally got the opportunity to get my hands back on the 600V bench supply. I need to repair my valve tracer, but firstly need the bench supply back again. Otherwise won’t be able to do all tests I want to around my 814 DC-coupled SE amplifier and many driver stages I want to try on my workbench before moving to the next stage.
An improved design to avoid fireworks
Tired of my old HT (+600V) variable bench power supply to suffer collateral damage when accidentally shorted whilst testing transmitting valves for output stage (i.e. FQP3n80c MOSFET passive regulator blowing out), I decided firstly to decide a simple and yet effective valve stabiliser. As nothing comes for free, these were my design constrain factors:
Input raw supply is +620V @ 100mA
Filament secondary winding is 15V @1.5A
No additional secondary winding is available for a floating screen supply (e.g. pass valve is pentode)
Output voltage ideally should be 0-600V
So with the restriction of not using a pentode as pass valve, I looked out for candidates to match my requirements and instantly thought about GU-50 in triode-strapped mode. Yes, I know that UG2 limit is 250V, not 1,000V as anode max voltage. But, in triode strapped specs are not shown. As recently checked this with the 814 triode strapped, and seems to be ok UG2=Ua in triode mode. 7N7 also said this was ok and Morgan Jones previously tested this as well with similar valves.
So, question here initially was: could the GU-50 withstand 600V in triode mode or should I needed to look out for other options?
After asking for some help in DIYaudio forum to see what was the best option on this topology and the recommendation was to use the GU-50 in “right-handed mode”:
GU50 right-handed
In this mode GU-50 was able to provide the regulation required (or close to it) with minimum driving requirements (i.e. 0V to +15V)
First version of the passive regulator:
The MOSFET is in source follower mode to provide the necessary grid current, albeit not sure how much grid current the GU-50 needed in right-handed mode. So the value of R6 must be adjusted on test. In order to survive an output short, R6 needs to be in the order of 100K and 3 or 4W to avoid blowing up the 18V zener protecting the grid.
The circuit above is very interesting as the anode dissipation of the GU-50 is very handy for this setup and requirements. Is important to highlight that the circuit has no regulation, so if this is a requirement a different topology must be explored.
If we look at the sandy option here, then the following equivalent is comparable:
R2 and R3 are required to protect silicon from output short. Q1 provides current limitation with R6. This topology suffers from same issues related to regulation as previous circuit.
A big issue on the two circuits presented before is that the potentiometer is stressed at full raw HT supply. This is far from ideal and despite the specific power requirements of the pot, we also need to ensure that the part can withstand the voltages used.
A slight modification (requiring an additional LT supply) can solve this problem:
Now P1 is connected to 15V. The gain of the M2 stage is significant so stability of circuit above is an issue now.
So if we have already introduced an LT supply in the circuit, then is a better choice to look at a feedback regulator.
A more complex circuit indeed, but a more effective one. The op-amp provides regulation by sensing output from R7 and R8 divider and comparing it to the stable reference from the output of P1. R10, R9 and C1 limit the HF gain response of the op-amp. R12/C3 and the 10pF FKP2 Wilma cap across R7 optimise the HF response of the overall regulator. D1 will protect the op-amp input (specially if 10pF cap is fitted). R1 is an additional protection for M1. M1 requires a good sink if wider regulation is needed. When output voltage is low and high current is drawn, then M1 is bearing all the effort and will dissipate a lot of heat (just do the maths).
The raw supply stays the same, however the gyrator stage was optimised as shown below:
The MOSFET was changed for am 1000V TO220FP plastic package one which is better from an insulation perspective given voltages used in this circuit. R5 and R6 changed to 150K 3W ones to provide protection to the zener and M1 in case output is shorted.
Previously I publish my 4P1L/4П1Л triode-strapped curves using my curve tracer. A forum friend who helped me in building my tracer as he was building his own, suggested using two tools for capturing curves (“Graph Grabber“) and plotting curves, loadlines, etc. (“Graph“).
Here is an example of the 4P1L/4П1Л triode-strapped curves using both applications to produce the safe operating area based on 7.5W anode dissipation. This can be extended to 9W if we consider the screen dissipation as well.
Continuing with the design of the 4-65A SE amplifier based on M. Koster design, I’m in the process of tweaking the 46 driver to optimise the operating point and provide maximum distortion to drive the demanding 4-65a. Here is the circuit I’m currently working on.
The current 46 driver will be biased at around 25-30mA using filament bias, so Vgk will be around -16 to -17.5V using a 10Ω filament bias resistor array. This will set the 46 at around 185-210V which will give sufficient headroom (i.e. need about 200Vpp max) to drive the 4-65a.
So today I look at varying slightly both anode currents and Vgk to see impact on THD.
So minimum THD is around -17V and Ia=30mA. So if setting the Rod Coleman filament regulator to ensure that Vgk=-17V and the anode gyrator to set anode voltage to ensure Ia is close to bias current would provide the minimum distortion (which is 0.04% in this example). Pa is close to 7W, but looking at the datasheet we can see that maximum Pa is 10W (as the latter 45 version).
46 Super Silvertone THD
So next I need to build a prototype of this driver with filament regulator and gyrator load.
Just got a couple of NOS EIMAC which I will be using in my SE design
4-65a EIMAC NOS
So did some test on distortion, transconductance and driving them to +22.22dBu output to check the quality of these two ones.
I used similar test rig as before. At some point will be able to get a proper filament supply for this valve, but for the time being I will continue to use the hum pot and the big electrolytic cap across my old bench power supply which can gently provide the 3.5A for the hungry filaments!
I tested them at the limit of my CCS and bench HT supply which at the moment cannot provide more than 360V @ 100mA.
Transconductance is in the region of 3,800 – 4,000 μmhos.
So biasing the valve at -2.5V and over 90mA of anode current, the harmonic profile looks like this:
You can test for leakage current using a simple amplifier made out of a NPN transistor and an indicator. In this case I used a Russian Neon (80V/0.5mA) and the existing supply on my tester (+/-80V). You can replace all this with a simple LED and the supply you have at hand. The circuit is designed to turn on the bulb when 5 μA leakage current is provided on the base of Q1 thanks to grounding the valve element next to the one under test. So for example if we want to measure cathode to grid leakage, we simply ground the cathode and we connect the tester to the grid. Same process is repeated with the other valve elements.
